Methods of operating an imaging pixel to accumulate charge from a photocurrent

ABSTRACT

An improved CMOS pixel with a combination of analog and digital readouts to provide a large pixel dynamic range without compromising low-light performance using a comparator to test the value of an accumulated charge at a series of exponentially increasing exposure times. The test is used to stop the integration of photocurrent once the accumulated analog voltage has reached a predetermined threshold. A one-bit output value of the test is read out of the pixel (digitally) at each of the exponentially increasing exposure periods. At the end of the integration period, the analog value stored on the integration capacitor is read out using conventional CMOS active pixel readout circuits.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application is a divisional of, and claims the benefit of,U.S. patent application Ser. No. 12/012,542, entitled “IMAGING PIXELCOMPRISING A COMPARATOR TO COMPARE INTEGRATED PHOTOCURRENT TO AREFERENCE VALUE AND DIGITAL OUTPUT CIRCUITRY,” filed Feb. 4, 2008, whichis a continuation of, and claims the benefit under 35 U.S.C. §120 of,U.S. patent application Ser. No. 11/427,483, entitled “MIXED ANALOG ANDDIGITAL PIXEL FOR HIGH DYNAMIC RANGE READOUT,” filed Jun. 29, 2006, theentire contents of both of which are incorporated herein by reference.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH

Not Applicable.

BACKGROUND OF THE INVENTION

The present invention is related generally to CMOS imaging sensors, andin particular to an improved CMOS imaging sensor having mixed analog anddigital pixel readout for high dynamic range.

Solid state image sensors (“imagers”) are important in a wide variety ofapplications including professional and consumer video and still imagephotography, remote surveillance for security and safety, astronomy andmachine vision. Imagers that are sensitive to non-visible radiation, forexample infrared radiation, are used in some other applicationsincluding night vision, camouflage detection, non-visible astronomy, artconservation, medical diagnosis, ice detection (as on roads andaircraft), and pharmaceutical manufacturing.

An image sensor comprises a two-dimensional array of photosensitiveelements (pixels) in combination with control and readout circuitry. Thepixels are sensitive to incoming radiation. The control and readoutcircuitry scans and quantitatively evaluates the outputs from the pixelsand processes them into an image.

FIG. 1 is a schematic block diagram and approximate physical layout of atypical conventional CMOS silicon imager. The imager comprises an n rowby an m column array of pixels implemented advantageously on a singlesilicon die. Each pixel contains a photo-detector plus control andmultiplexing circuitry. An active pixel is a pixel which also includessignal amplification and processing circuitry. Each pixel generates anoutput signal that is proportional to the accumulated radiation incidenton the photo-detector during a defined integration period.

All the pixels in a single row are controlled by a set of row signalsgenerated by a row multiplexer. The row multiplexer contains circuitsthat perform row address and timing functions within the pixel includingpixel reset and controlling the length of the integration period. Allpixels in a single row output onto their respective column bus at thesame time, but pixels in different rows can output at different times.This staggering allows the pixels in a column to share the column bus,multiplexing their output signals sequentially onto the column bus onerow at a time.

All the pixels in a single column send their output signals to a columnmultiplexer via the column bus. The pixel output signals are multiplexedonto the column bus in response to control signals from the rowmultiplexer. Circuits within the column multiplexer can perform a numberof functions including amplification, noise reduction and multiplexinginto predefined video or image formats, e.g. a standard TV videosequence. The video or image signals generated by the column multiplexercan be further processed by an on-chip image signal processor toreorganize, improve and enhance the image.

FIG. 2 is a circuit schematic of a typical conventional CMOS activepixel, commonly known as a 3-T cell. The pixel comprises aphoto-detector, an integration capacitor C_(int), a source followerdevice M1, a pre-charge device M2, and a row-select device M3. Theintegration capacitor may simply be the parasitic capacitance of thephoto-detector and M1. The active pixel is controlled by two rowsignals, pre-charge and row-select. It also connects to the columnoutput bus which is terminated in the column multiplexer with a currentsource or other suitable load device.

At the beginning of an integration cycle, a pulse on the pre-charge linecharges the integration capacitor to a known value via M2. During theintegration period, photocurrent generated by the photo-detector inresponse to incident radiation discharges the integration capacitor.This causes the voltage V_(s) on the gate of M1 to change. The change involtage ΔV_(s) is a function of the accumulated photo-charge ΔQ and theintegration capacitance C_(int) according to ΔV_(s)=ΔQ/C_(int). Theratio of output voltage to accumulated charge ΔV_(s)/ΔQ=1/C_(int) isknown as the conversion gain. At the end of the integration period, therow-select line is set to allow the voltage V_(s) to be read out on tothe column output bus via M1 and M3. The operation of this type of pixelis well understood by those skilled in the art.

For a given integration period, the minimum optical signal that a pixelcan detect is limited by shot noise in the photo-detector, reset noisein the integration capacitor (sometimes known as kTC noise) andelectrical noise in the read-out circuitry. The maximum optical signalthat a pixel can detect is limited by the charge accumulation capacityof the integration capacitor. Once this limit is reached, the pixel issaid to be saturated. The dynamic range of a pixel, typically measuredin dB, is the ratio of the maximum optical signal (at saturation) to theminimum optical signal (limited by noise). The dynamic range of a pixelis a measure of the imager's ability to capture both very bright andvery dark objects in a single image.

The dynamic range of an imager is the ratio of the maximum opticalsignal that can detected without pixel saturation to the minimum opticalsignal that can be detected, allowing for changes in integration timeand aperture. The dynamic range of an imager can be much larger than thedynamic range of the pixel. Note, however, that modifications tooperation of the imager such as integration time and aperture affect allof the pixels in an imager equally. They allow an imager to operate invery bright light or under very low light conditions. They do not,however, improve an imager's ability to capture very bright and verydark objects in the same scene.

There are a number of applications of CMOS imagers that require veryhigh dynamic range within a single scene. An example is an automotivenight vision camera in which a scene to be processed may include bothvery dark objects (e.g. animal or pedestrian on the road at night) andvery bright objects (oncoming car headlamps). Another example is asecurity camera which is used to identify a poorly lit person against abright sunlit background. These applications require an imager in whichthe pixel has a very high dynamic range (e.g. 100 dB). The pixels usedin CMOS cameras (for example, the pixel shown in FIG. 2) typically havea pixel dynamic range of 70 dB or less. They are not suitable,therefore, for these high dynamic range applications. For theseapplications, a pixel with increased dynamic range is required.

An obvious approach to increasing the dynamic range of a pixel is toincrease the value of the integration capacitor. This increases thesaturation level of the pixel. Unfortunately, it also reduces theconversion gain of the pixel which reduces sensitivity of the imager andtherefore the signal to noise ratio at low light levels. So the netimprovement in dynamic range may be small. Increasing the size of theintegration capacitor also causes the area of the pixel to increasesignificantly to accommodate the extra capacitance.

One technique that has been proposed for increasing the dynamic range ofpixels is to use a non-linear element to compress the output of thepixel. The photodetector current may, for example, be fed into alogarithmic current to voltage converter such as a diode-connected MOStransistor. While such devices can achieve very high dynamic range, theysuffer from poor sensitivity, low signal to noise ratio and exhibit highlevels of fixed pattern noise.

A second technique uses conventional charge accumulation when theillumination level is low but records the ‘time to saturation’ underhigh levels of illumination. Once a nominal saturation level isachieved, a comparator switches to sample the voltage of an analog rampthat is supplied to every pixel. The sampled voltage provides a measureof the time instant when saturation occurred. This scheme requires a lownoise analog ramp and precision components within the pixel to achievelow noise. It also suffers from high power dissipation because theprecision comparator is “always on”.

A third technique uses an overflow gate to dynamically adjust thesaturation level during integration. In a conventional 3-T pixel, thegate of the pre-charge device is pulsed “high” at the beginning of theintegration cycle and then held “low” during the integration period. Byplacing a small positive control voltage on the gate of the pre-chargedevice during the integration period, one can effectively lower thesaturation level of the pixel. During the initial portion of theintegration period, the saturation level is set low. Once saturation isreached, any additional photocurrent is drained away through thepre-charge device. After a predetermined interval, the saturation levelis raised. Charge can then once again accumulate until the newsaturation level is reached. The saturation level is monotonicallyincreased in steps during the integration period in such as way as tocreate a well defined non-linear charge to voltage relationship. Thisscheme enhances dynamic range at the expense of reduced signal to noiseratio due to an effectively reduced charge storage capacity.

A fourth technique proposes a reset gate to drain optical charge fromthe photo-detector. By selectively activating the reset gates ofindividual pixels, one can individually set the effective integrationtime of each pixel. This technique was proposed for use with CCDimagers. It could also be applied to CMOS imagers. However, control ofthe reset gate is external to the pixel array. It thus requiresconsiderable external circuitry to remember the recent activity of eachpixel and then a complex 2-D addressing scheme within the array toindividually control the reset gate of each pixel.

It has further been suggested to employ a multiple sampling techniquebased on pixel level analog-to-digital (A/D) conversion. The in-pixelA/D converter uses a technique known as multi-channel bit-serial (MCBS)to convert the analog output of the pixel to a Gray code digital output.A block diagram of the A/D converter is shown in FIG. 3. It comprises acomparator, and a D-latch. This circuit generates one bit of the Graycode digital output Dout at a time. The voltage to be converted Ain issupplied to one input of the comparator. An analog ramp Aramp issupplied to the other input of the comparator. An m bit Gray codeddigital ramp Dramp, whose digital value at any time t corresponds to thevalue of the analog ramp at time t is also provided. The i^(th) bit ofthe digital output Dout_(i) is determined by supplying Dramp_(i) to theD input of the latch. Dramp_(i) is a binary digital waveform whose valueat any time t is equal to the i^(th) bit of the digital Gray code ramp.When Aramp is equal to the input value Ain, the comparator switches andstores the appropriate digital value into the latch. This process isperformed for each of the m bits of the digital output. Gray code isused so as to minimize errors that would be caused by small changes inthe input while generating a multi-bit digital output.

The output of the pixel is sampled by the A/D converter at a series of kexponentially increasing integration times T, 2T, 4T, . . . , 2 ^(k)T.The digitized output of the A/D converter will approximately double eachtime until saturation is reached. Suppose that saturation is firstdetected when performing the j^(th) conversion, that is, after time2^(j)T. The output of the pixel is then the m bit output of the A/Dconverter after the (j−1)^(th) conversion (last output prior tosaturation) multiplied by a scale factor 2^(k−j+1) to account for thereduced integration time. This scheme has the advantage that all A/Dconversion is performed inside the pixel; no analog output is required.It also increases the dynamic range of the pixel by a factor of 2^(k)while providing m bits of resolution at all levels of illumination.

With this approach, it is not necessary to output all m bits at eachsampling instant. All m bits are output at the first sampling instant(at time T). Assuming the charge to voltage response of the pixel islinear, the output at time 2T will be double the output at time T. Oncethe scaling factor is applied, the outputs will be identical except forthe least significant bit (LSB). This LSB is the one bit of addedprecision that has been obtained by doubling the integration time. Afterthe first integration time, therefore, it is only necessary to outputthe LSB of the m bit digital output. Each subsequent sampling instantyields another bit of the digital output until saturation is reached.

There are, however, a number of problems associated with this scheme.Firstly, the accuracy of the digital readout requires that the opticalillumination be constant over the integration period and that theconversion from light to charge and charge to voltage be linear.Variations in illumination and/or non-linearities in the circuitinvalidate the assumption that the output at time 2T will be double theoutput at time T and hence the property that only the LSB of theconverted output will change after the first sample. Smallnon-linearities, for example, can cause inconsistencies between theoutput bits in successive samples which can, in turn, cause (potentiallylarge) errors in the digital readout. Secondly, the sensitivity of thepixel is limited by the resolution of the MCBS converter which is, inturn, limited by the gain-bandwidth product of the comparator and theaccuracy of the analog and digital ramps. Area and power limitationswithin the pixel preclude the use of a high-gain, high-bandwidthcomparator. Quantization noise in the converter is therefore likely tobe much greater than kTC noise or the analog read-out noise that wouldbe found in a conventional imager. This will likely limit the use ofthis type of pixel in low-light conditions.

Accordingly, it would be advantageous to provide a CMOS imaging sensorwith a pixel design which has a large dynamic range without compromisinglow-light performance.

BRIEF SUMMARY OF THE INVENTION

Briefly stated, the present invention provides an improved CMOS pixelwith a combination of analog and digital readouts to provide a largepixel dynamic range without compromising low-light performance. Thepixel design uses a comparator to test the value of an accumulatedcharge at a series of exponentially increasing exposure time intervals.A one-bit output value of the test is read out of the pixel (digitally)at each of the exponentially increasing exposure periods, with a firstvalue representing an accumulated charge below a predeterminedthreshold, and a second value representing an accumulated charge equalto or exceeding the predetermined threshold. At the end of theintegration period, the analog value stored on the integration capacitorat the end of the time interval in which the accumulated charge equaledor exceeded the predetermined threshold is read out using conventionalCMOS active pixel readout circuits.

In one embodiment, the improved CMOS pixel includes a photo-detector, anintegration capacitor, a source follower device, a pre-charge device,and a row-select device. In addition, the CMOS pixel includes a clockedcomparator, an RS flip-flop, a photocurrent switch device, and a digitalrow-select device. The CMOS pixel is controlled by four row signals:pre-charge, sample, analog-row-select and digital-row-select, and isconnected to two column output lines: the analog column bus and thedigital column bus.

The foregoing features, and advantages of the invention as well aspresently preferred embodiments thereof will become more apparent fromthe reading of the following description in connection with theaccompanying drawings.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

In the accompanying drawings which form part of the specification:

FIG. 1 is a schematic block diagram of a prior art conventional CMOSsilicon imager;

FIG. 2 is a circuit schematic of a prior art CMOS active pixel;

FIG. 3 is a block diagram of a prior art CMOS pixel A/D converter;

FIG. 4 is a block diagram of an embodiment of the CMOS pixel of thepresent invention;

FIG. 5 is a timing diagram showing an example of the operation of theCMOS pixel of the present invention; and

FIG. 6 is a circuit schematic of an embodiment of the CMOS pixel of thepresent invention.

Corresponding reference numerals indicate corresponding parts throughoutthe several figures of the drawings. It is to be understood that thedrawings are for illustrating the concepts of the invention and are notto scale.

DESCRIPTION OF THE PREFERRED EMBODIMENT

The following detailed description illustrates the invention by way ofexample and not by way of limitation. The description enables oneskilled in the art to make and use the invention, and describes severalembodiments, adaptations, variations, alternatives, and uses of theinvention, including what is presently believed to be the best mode ofcarrying out the invention.

In general, an improved CMOS pixel of the present invention utilizes acombination of analog and digital readout to provide a large pixeldynamic range without compromising low-light performance using acomparator to test the value of accumulated charge at a series ofexponentially increasing exposure times. Analog to digital conversion isnot performed within the pixel. The test is used to stop the integrationof photocurrent once the accumulated analog voltage has reached apredetermined threshold. The one-bit output value of the test is readout of the pixel (digitally) at each of the exponentially increasingexposure periods. At the end of the integration period, however, theanalog value stored on the integration capacitor is read out usingconventional CMOS active pixel readout circuits. The accuracy of theoutput is not a function of the precision of the comparator and there isno quantization noise to limit low-light performance.

A block diagram of an embodiment of the pixel of the present inventionis shown generally at 100 in FIG. 4. Like the 3-T pixel, the embodimentof the pixel of the present invention comprises a photo-detector 102, anintegration capacitor C_(int), a source follower device M1, a pre-chargedevice M2, and a row-select device M3. In addition, it includes aclocked comparator 104, an RS flip-flop FF, a photocurrent switch deviceM4, and a digital row-select device M5. The pixel 100 is controlled byfour row signals: pre-charge, sample, analog-row-select, anddigital-row-select. It also connects to two column output lines: theanalog column bus and the digital column bus.

At the beginning of an integration period for the pixel 100, thepre-charge signal sets RS flip-flop FF, enabling pass transistor M4. Asin a conventional active CMOS pixel, the pre-charge signal also chargesnodes dnode and inode via M2 and M4. Once the pre-charge signal isreleased, photocurrent begins to discharge inode via M4. At well definedexponentially increasing sub-integration times T, 2T, 4T, . . . etc.,the signal sample triggers the comparator 104 to compare the accumulatedsignal at inode to a predetermined threshold Vref. Vref is chosen sothat the RS flip-flop FF will reset when the integrating node inode hasreached not quite 50% of its overall charge holding capacity. After eachcomparison event, the digital-row-select signal is pulsed. This causesthe output of RS flip-flop FF to be read out over the digital column busvia M5. Once RS flip-flop FF has been reset by one of these comparisonevents, M4 is turned off. This isolates the photodiode from inode,thereby stopping the integration of photocurrent. At the end of thetotal integration period, the accumulated analog charge on inode is readout over the analog column bus via M3 in the conventional manner.

The output of each pixel 100 is thus a series of single bit digitaloutputs (one for each sub-integration period) plus an analog output. Theintegration of photocurrent within the pixel 100 is stopped whenever theaccumulated charge is more than 50% of capacity. Since eachsub-integration period is double the previous period, this ensures thatthe integrated signal on inode never exceeds 100% of capacity (unlessthe optical signal is so strong that it saturates the pixel in the firstsub-integration period T). The digital output sequence records whenintegration was stopped, and may be used to generate what is effectivelyan exponent qualifying the analog output (mantissa).

For example, assume that a pixel has an analog output range of 0(precharge) to 1.0 (saturation) and that the total integration period is32T. The accumulated charge is tested at times T, 2T, 4T, 8T and 16T asshown in FIG. 5. Further assume that the threshold voltage is set tostop integration when the accumulated charge reaches 40% of capacity,and that the optical signal is such that it would normally saturate thepixel 100 in time 6T. At the end of the first sub-integration period T,the pixel 100 will be at 17% of capacity, so the comparator does notfire and the RS flip-flop FF remains set. The digital output is 1. Atthe end of the second sub-integration period 2T, the pixel 100 is at 33%of capacity, so the comparator does not fire and the RS flip-flop FFremains set. The digital output is again 1. At the end of the thirdintegration period 4T, the pixel is at 67% of capacity, so thecomparator does fire and resets the RS flip-flop FF which stops theintegration. The digital output for this and all remainingsub-integration periods is 0.

At the end of the total integration period the analog output 0.67 isread out. The digital output sequence is 11000. This indicates that theanalog output of 0.67 was accumulated in a time period of 4T. The scaledpixel output is thus (32÷4)×0.67, which equals 5.33. This represents thesignal that would have been accumulated in a time period of 32T if theintegration capacity were not limited. Note that the maximum signal thatcan be measured in this example is one that just saturates the pixel intime period T. This example represents a 32 times increase in thedynamic range of the pixel 100 compared to a 3-T pixel with the samecharge accumulation capacity. For optical signals not high enough toswitch the comparator at any point during the total integration period,the SNR remains the same as the original 3-T pixel. For optical signalshigh enough to switch the comparator, the final analog signal willalways lie somewhere between 40% and 80% of capacity. Maximum SNR hastherefore only been reduced by 20%.

The technique of the present invention increases dynamic range of thepixel 100 without significantly reducing the maximum SNR, and does notperform in-pixel A/D conversion. The accuracy is not limited by theprecision and speed of the comparator 104 nor the threshold voltage.Low-light performance is limited only by shot noise and analog read-outnoise. There will be kTC noise associated with the reset of inode, butthis can be removed using external digital correlated doubled samplingas with a conventional active pixel. In contrast, kTC noise (as opposedto offset errors) cannot be removed from prior art pixel designsincorporating in-pixel analog-to-digital conversion because it is toosmall to be detected by the in-pixel A/D technique.

The technique of the present invention assumes that the charge tovoltage versus time process is linear, and small non-linearities do notproduce large errors. The technique does not depend on an arithmeticrelationship between the values of the digital output bits. The outputbits simply define when integration was stopped. The output will alwaysbe a sequence of ones followed by a sequence of zeros—giving anunambiguous measure of integration period. Similar to a conventionalactive CMOS pixel, the accuracy of the pixel 100 output is purely afunction of the linearity and accuracy of the analog output. The digitalsignals merely scale the output by capturing the value of the opticalintegration period.

The photocurrent switch M4 will add charge-feed through and kTC noise tothe final output. Under low light conditions, the comparator 104 willnever fire and so M4 remains conducting. The presence of. M4 does not,therefore affect the low light performance of the pixel. Charge-feedthrough and kTC noise due to M4 will only occur under conditions wheninode is at least 50% of capacity, at which point they will notsignificantly affect the signal to noise ratio.

A challenge in implementing this new pixel is to provide thefunctionality of FIG. 4 without significantly increasing the complexityof the pixel circuitry which would, in turn, significantly reduce thefill factor. FIG. 6 shows one possible CMOS implementation of the pixel100 of the present invention. Those of ordinary skill in the art willrecognize that other circuit configurations which accomplish the samefunctionality may be implemented without departing from the scope of theinvention, and may have lesser or greater fill factors.

As in the 3-T pixel, the new pixel 100 comprises a photo-detector 102,an integration capacitor C_(int), a source follower device M1, apre-charge device M2, and an analog row-select device M3. M4 and M6function as a complimentary CMOS switch controlled by signals full andfull which disconnect the photodiode from the integration node inodeonce the pixel 100 is deemed full.

The device pairs M10/M11 and M12/M13 form two inverters which act as acomparator within the pixel 100. The switching voltage of the comparator(Vref in FIG. 4) is simply the switching voltage of inverter M10/M11which is determined by the relative sizes of devices M10 and M11. Thesewill, of course, vary according to process variations across the imager.As pointed out previously, however, the correct operation of the pixel100 does not depend on the exact switching voltage of the comparator.

A second source follower device M8 is terminated by a switched currentsource load device M9. When M9 is conducting, the output of M8, test, isa replica of the analog output voltage that will be produced by theregular source follower device M1. The signal test connects to the inputof the comparator.

At the beginning of the integration cycle, the input sample is set low,turning off device M9. The pre-charge signal pch is taken low, causingdnode, inode, and the signal full to be all set high. This, in turn,causes the signal test to be set above the comparator threshold whichalso drives full high. Once pch is taken high, the comparator latchesinto the “not full” state due to the positive feedback supplied throughdevice M8.

At well defined exponentially increasing sub-integration times T, 2T,4T, . . . etc., the signal sample momentarily switches to a smallpositive voltage, sufficient to cause M9 to act as a current source loadto M8. The signal test will then be equal to the signal inode minus thethreshold voltage of M8. If little optical charge has accumulated oninode, the voltage of test will be above the comparator threshold andthe comparator/latch formed by the devices M10 through M13 will remainin their preset state.

However, if the node inode has accumulated sufficient charge to be near50% of its capacity, the voltage at test will be low enough to causeM10/M11 to switch, causing full to go high. This will, in turn, causeM12/M13 to switch which will cause full to go low. This will disconnectthe photodetector 102 from inode by turning off M4 and M6, therebypreventing any further accumulation of optical charge. When full goeslow, it also lowers the drain of M8 latching the comparator into thefull state. Once this has occurred, the comparator/latch will not resetinto the full state until a pch pulse is applied at the beginning of thenext integration cycle.

A digital row select signal is supplied to read the digital output ofthe comparator/latch via switch device M5 after each sample event. Atthe end of the integration period, the analog row select line causes theaccumulated analog charge on inode to be read out in the usual fashionvia switch device M3.

Note that for most of the integration period, the signal sample is heldlow. This is to reduce power dissipation in the pixel 100 by turning offload device M9. It is only pulsed to a small positive voltage when acomparison needs to be made.

As various changes could be made in the above constructions withoutdeparting from the scope of the invention, it is intended that allmatter contained in the above description or shown in the accompanyingdrawings shall be interpreted as illustrative and not in a limitingsense.

The invention claimed is:
 1. A method comprising: generating a photocurrent representative of radiation incident on a photodetector of a pixel during a time period; accumulating on a capacitor a charge at least partially from the photocurrent; producing a first output signal of the pixel representing the charge accumulated on the capacitor; and producing a second output signal of the pixel indicative of an amount of time required for the charge accumulated on the capacitor to reach a reference level, wherein the second output signal is produced using an RS flip-flop circuit, wherein the first output signal is an analog signal, wherein the second output signal is a digital signal, wherein the method further comprises modifying the first output signal at an end of the time period using the second output signal to produce a resulting signal representing an amount of the radiation incident on the photodetector, wherein producing the second output signal comprises comparing, at one or more times during the time period, the charge accumulated on the capacitor to the reference level; and (a) responsive to the charge accumulated on the capacitor being less than the reference level, generating a first digital value; and (b) responsive to the charge accumulated on the capacitor exceeding the reference level, generating a second digital value.
 2. The method of claim 1, further comprising, responsive to the charge accumulated on the capacitor exceeding the reference level, holding the first output signal at a current level.
 3. The method of claim 2, wherein holding the first output signal at a current level comprises disconnecting the capacitor from the photodetector to stop accumulating charge on the capacitor from the photocurrent.
 4. The method of claim 3, wherein the digital signal is a one-bit digital signal.
 5. The method of claim 3, wherein modifying the first output signal at the end of the time period using the second output signal comprises generating an exponent qualifying the first output signal at the end of the time period using the second output signal.
 6. The method of claim 3, wherein the second output signal comprises a sequence of digital values produced by the comparison, at the one or more times during the time period, of the charge accumulated on the capacitor to the reference level, each digital value of the sequence of digital values having a position within the sequence identifying at which time of the one or more times during the time period the digital value was produced.
 7. A method comprising: generating a photocurrent representative of radiation incident on a photodetector of a pixel during a time period; accumulating on a capacitor a charge at least partially from the photocurrent; producing a first output signal of the pixel representing the charge accumulated on the capacitor, wherein the first output signal is an analog signal; and producing a second output signal of the pixel indicative of an amount of time required for the charge accumulated on the capacitor to reach a reference level, wherein the second output signal is a digital signal, and wherein producing the second output signal comprises comparing, at one or more times during the time period, the charge accumulated on the capacitor to the reference level; and (a) responsive to the charge accumulated on the capacitor being less than the reference level, generating a first digital value; and (b) responsive to the charge accumulated on the capacitor exceeding the reference level, generating a second digital value and holding the first output signal at a current level, wherein holding the first output signal at a current level comprises disconnecting the capacitor from the photodetector to stop accumulating charge on the capacitor from the photocurrent; wherein the second output signal comprises a sequence of digital values produced by the comparison, at the one or more times during the time period, of the charge accumulated on the capacitor to the reference level, each digital value of the sequence of digital values having a position within the sequence identifying at which time of the one or more times during the time period the digital value was produced; and wherein the method further comprises modifying the first output signal at an end of the time period using the second output signal to produce a resulting signal representing an amount of the radiation incident on the photodetector, wherein modifying the first output signal at the end of the time period using the second output signal comprises generating an exponent qualifying the first output signal at the end of the time period using the second output signal, wherein generating the exponent comprises generating the exponent from the sequence of digital values.
 8. The method of claim 7, wherein the one or more times during the time period define one or more intervals of the time period, and wherein the sequence of digital values identifies a discrete interval of the one or more intervals at which the charge accumulated on the capacitor exceeds the reference level.
 9. The method of claim 8, wherein the one or more intervals comprises a first interval and one or more additional intervals, each of the one or more additional intervals having a duration exponentially greater than a duration of an immediately preceding interval.
 10. The method of claim 9, wherein the reference level is selected such that if the charge exceeds the reference level during an interval of the time period, the first output signal will remain below a maximum limit during a next consecutive interval of the time period. 